Wireless communication devices, such as cellular handsets and wireless personal digital assistants, continue to gain widespread use in today's world. With each new generation of wireless communication technology, they have become more technically sophisticated, now commonly providing or supporting, in addition to traditional voice communications, text messaging, electronic mail, mini browsers for surfing the Internet, etc. While users no doubt enjoy the benefits these additional features provide, the additional features can seriously impact battery life.
Efforts to extend battery life focus in large part on ways to improve the energy efficiency of the wireless communication device's RF transmitter, particularly the RF transmitter's power amplifier (RFPA), since the RFPA is usually the dominant energy consumer. Unfortunately, these efforts are complicated by the fact that many wireless communication systems employ nonconstant-envelope modulation schemes. Nonconstant-envelope modulation schemes are used to increase spectral efficiency, i.e., to increase the amount of information that can be transmitted in a given bandwidth of the RF spectrum. However, their use also hinders the ability to increase the energy efficiency of conventional RF transmitters (e.g., quadrature modulator based RF transmitters).
When nonconstant-envelope modulation schemes are used in conventional RF transmitters, the output power of the RF transmitter's RFPA must be backed off to prevent the RFPA from clipping the nonconstant-envelope signals. Maintaining linearity also requires that the RFPA be configured to operate exclusively in its linear region of operation (i.e., that a ‘linear RFPA’ be used). Failure to back off the output power and use a linear RFPA results in signal distortion at the output of the RFPA. The signal distortion makes it difficult to comply with noise limitation specifications imposed by communications standards.
The need to back off output power and use a linear RFPA undesirably results in a sacrifice of energy efficiency for linearity. Fortunately, an alternative type of transmitter, commonly known as a polar transmitter, is available. A polar transmitter avoids the efficiency versus linearity trade-off by temporarily removing the amplitude modulation from the nonconstant-envelope signal. The constant-envelope signal that remains contains only phase modulation. It is translated to RF to create a constant-envelope phase-modulated RF signal and then amplified by the RF transmitter's RFPA. Because the constant-envelope phase-modulated RF signal has a constant envelope, there is no risk of signal peak clipping and the polar transmitter's RFPA can be implemented as a highly efficient nonlinear RFPA.
As the phase-modulated constant-envelope RF signal is amplified by the nonlinear RFPA, the power supply port of the nonlinear RFPA is modulated by a time-varying (i.e., dynamic) power supply containing the signal envelope. In effect, the nonlinear RFPA operates as an amplitude modulator, superimposing the signal envelope onto the phase-modulated constant-envelope RF signal, to form the desired amplitude- and phase-modulated RF output signal at the RF output of the nonlinear RFPA.
To maximize efficiency, the nonlinear RFPA is usually implemented as a switch-mode RFPA. As shown in FIG. 1, a switch-mode RFPA 100 comprises a transistor 102, typically a bipolar transistor of some sort (although field-effect transistors (FETs) are alternatively used), an RF choke 104, a bias resistor 106, and a matching network 108. The base of the transistor 102 is configured to receive the phase-modulated constant-envelope RF signal; the collector is configured to receive a dynamic power supply voltage VPA containing the signal envelope (via the RF choke 104); and the emitter is coupled to ground via the bias resistor 106. The RF choke 104 is used to prevent RF energy from filtering into the power supply circuitry used to generate the dynamic power supply voltage VPA. The matching network 108 operates to block harmonic components from reaching the load 110 (the antenna of the polar transmitter) and to shape the amplitude- and phase-modulated RF signal to its desired signal characteristics.
The transistor 102 of the switch-mode RFPA 100 can be viewed as a switch, as illustrated in FIG. 2. The transistor 102 is switched ON (closed) and OFF (open), between saturation and cut-off, in response to the constant-envelope phase-modulated RF signal applied to the base of the transistor 102. Ideally, when the transistor 102 is ON, no voltage is dropped across the collector-emitter terminals of the transistor 102, and when the transistor 102 is OFF, no collector current IC flows through the transistor 102. Hence, at least in theory, the collector-emitter voltage VCE and collector current IC waveforms do not overlap in time and the switch-mode RFPA 100 achieves very high efficiency.
The transistor 102 of the switch-mode RFPA 100 typically comprises a bipolar transistor, such as a heterojunction bipolar transistor (HBT). Bipolar transistors exhibit a nonzero collector-emitter saturation voltage VCE,SAT when the transistor 102 is in the ON state, as shown in the characteristic curves of a typical bipolar transistor in FIG. 3. Note that in the ON state, the collector current IC and collector-emitter saturation voltage VCE,SAT are both nonzero. Further, in the OFF state the transistor 102 still passes a small amount of collector current (i.e., the transistor 102 does not serve as a completely open switch). Accordingly, unlike the ideal case discussed above, the collector current IC and collector-emitter voltage waveforms do overlap to some extent, both in the ON and OFF states, and the switch-mode RFPA 100 does not achieve the theoretical 100% efficiency. Despite the reduction in efficiency, the switch-mode RFPA 100 is still significantly more efficient than linear RFPAs.
Although the reduction in efficiency of the switch-mode RFPA 100 due to the nonzero collector-emitter saturation voltage VCE,SAT may be tolerable in many applications, the presence of the collector-emitter saturation voltage VCE,SAT can cause other problems that may not be tolerable. One particularly significant problem relates to the DC offset voltage that forms at the collector of the transistor 102 due to the nonzero collector-emitter saturation voltage VCE,SAT of the transistor 102 in the ON state. For example, when the switch-mode RFPA 100 is employed in a polar transmitter, the DC offset voltage can cause envelope inaccuracies in the signal envelope of the amplitude- and phase modulated RF output signal. This problem is further complicated by the fact that the DC offset voltage tends to drift with temperature, usually varies from part-to-part, and changes as the switch-mode RFPA ages. DC offset voltage drift (or “DC offset drift” for short) is highly undesirable since the distortion it causes to the signal envelope of the envelope of the amplitude- and phase modulated RF output signal results in a reduction in modulation accuracy and an increase in spectral regrowth into adjacent communications channels. The effects on the modulation accuracy and the increase in spectral regrowth can be severe enough that compliance with communications standards is difficult or impossible to achieve.
Various approaches to compensating for DC offset drift have been proposed. However, they have a number of drawbacks. One approach employs a thermal probe and a behavioral model determined in the laboratory to determine a temperature dependent voltage function, which is then used to compensate for DC offset drift. However, in addition to the drawback of having to use a thermal probe, that approach does not adequately account for part-to-part variations and requires multiple temperature measurements and/or expensive sorting procedures, which are not well suited for a mass production environment. Other approaches not requiring the use of a thermal probe have been proposed. However, those approaches require sophisticated and expensive wideband power detectors capable of providing a continuous and linear response over a wide dynamic range, as well as complex filtering mechanisms to prevent RF energy from the switch-mode RFPA from interfering with other circuitry of the transmitter.
It would be desirable, therefore, to have methods and apparatus for dynamically compensating for DC offset drift and other process, voltage, and temperature related signal variations in switch-mode power amplifiers (PAs) and in polar transmitters employing switch-mode PAs, which avoid the drawbacks and limitations of conventional DC offset drift compensation approaches.